Hybrid power amplifier circuits. Hybrid linear HF power amplifier

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When repeating similar “hybrid” power amplifiers, many radio amateurs are faced with the problem that the power amplifier with two GI-7B lamps proposed, for example, by S. Voskoboynikov (UA9KG) “does not deliver” the required 600 W. Let's try to understand the examples and mistakes that a large number of radio amateurs make in the article below.

The desire to express my thoughts on such, in general, not a new topic as a hybrid power amplifier, appeared after reading the article and based on my experience. The performance characteristics given by the author of this article, unfortunately, are not achievable. In particular, the output power of this stage, in the version in which it is published, will not exceed 360W. To get such power from two GI-7B lamps is, to put it mildly, irrational. So why doesn’t this cascade “deliver” the 600 W promised by the author? Let us briefly consider the operation of this cascade, Fig. 1.

To begin with, it should be recalled that GI-7B lamps, by the way, like most metal-ceramic microwave triodes, are lamps with an “average” anode-grid characteristic. To obtain a quiescent current of 30...40 mA to the lamp, with an operating anode voltage of about 2 kV, it is necessary to apply a negative bias to the grid - 25 V or, which is the same thing, to give a positive potential to the cathode by the same amount. The excitation voltage applied to the base of transistor VT1 opens it with a positive half-wave. The voltage at the collector and, accordingly, at the cathode of the lamp decreases, as a result of which the current through the lamp increases.

The negative half-wave closes the transistor, the voltage on the collector increases, the current through the lamp decreases, because the potential difference between the cathode and grid section increases. From the point of view of the cascade energy, we are only interested in the positive half-wave of the exciting voltage, due to the fact that the negative half-wave, when idealizing the input characteristics of the lamp, does not cause anode current and lies in the cutoff region.

The conclusion suggests itself: the amplitude of the RF voltage on the collector, and it is precisely this that is the exciting voltage for the lamp, lies between two boundary conditions. Below is the saturation voltage at the collector (or cathode) at the resting point, about 25V.

From this it is clear that the amplitude of the HF voltage at the cathode of the lamp is equal to:

(1) U to ex. = U n k-e - U k-e us.

Voltage U k-e us. depending on the type of transistor is 0.5...2.5V. In practice, it should be chosen at least 5V, since at lower voltages on the collector, the amplifying properties of the transistor tend to zero. The value of U k-e us. there is a voltage at the collector (cathode) for a given quiescent current in a circuit with a galvanically grounded grid.

In our example, U n k-e is 25V. In general, this value is taken from the input characteristics of the lamp. Substituting these values into formula (1) we obtain U to exc - 20 V. Next, it is not difficult to calculate the power supplied by the cascade. Anode current pulse amplitude:

(2) I to A max. = U to exc x S = 2 0 x 46 = 0.92 A, where:

S is the total slope of the characteristics of two lamps.

Constant component of the anode current:

(3) I ao = I a max x K o = 0.92 x 0.33 = 0.3A. where Ko = 0.33 is the expansion coefficient of the cosine pulse for a cutoff angle of 90 degrees (class B) and taking into account the quiescent current of the lamp.

Power supplied to the anode circuit of the lamp, U a = 2 kV:

(4) P sub = I a o xU a = 0.3 x 2000 = 600 W.

Assuming the efficiency of the cascade is about 60%, we obtain the power in the load P n = P in x efficiency = 600 x 0.6 = 360 W.

It is clear that the resulting power in the load is unlikely to satisfy. How to increase power? After all, the same lamps, in a classic design with a common grid, deliver up to 1 kW to the load. From the analysis of the circuit it can be understood that the main parameter limiting the power is the excitation voltage U exc. which in turn is related to the lamp bias voltage.

It is clear that the transistor operates in a very irrational mode, with collector power. You can increase this voltage by reducing the bias at the base of the transistor, but then the quiescent current will decrease unacceptably and the cascade will go into mode C. This is where we come to the main idea. Let's consider a slightly modified version of the scheme, Fig. 2.

Fig.2.

As you can see, the scheme is almost the same. Unless a positive (!) bias is applied to the grid, and at HF it is grounded through blocking capacitors C bl.

What has changed for the lamp? Absolutely nothing. After all, in order to obtain the same quiescent current, the potential difference in the cathode-grid section must remain the same. It remained so, however, the potentials of the cathode and grid relative to the common wire increased by the value Ucm. But for the transistor, very significant changes have occurred. The voltage at its collector increased by the value Ucm. and it became:

(5) U" k-e = U k-e - U cm., where:

U k-e - voltage for the circuit in Fig. 1.

In other words; we managed to increase the voltage on the collector (cathode) without changing the quiescent current of the lamp. Now we can consider a more complete circuit diagram of the output stage, Fig. 3.

Fig.3.

Resistor R1 (in the grid circuit) is not involved in the operation of the cascade and is intended to provide galvanic connection to ground in receive mode. The ratings of the basic divider R3...R5 are not indicated, because voltage on the TX bus??? different in different designs.

To ensure normal thermal stabilization of the operating point, the current flowing through the divider must be at least

(0.01...0.15) * I to max. = 100 mA.

A few words about the choice of the value of U cm. It cannot be raised indefinitely, since at a constant quiescent current the voltage U" k-e also increases. This value can be determined from the inequality:

U cm.< U n k-e add. - U k-e, where:

U n k-e add.- maximum permissible voltage on the collector (reference value);

U k-e - voltage on the collector for a given quiescent current in a circuit with a galvanically grounded grid (from the input characteristics of the lamp).

The zener diode protects the transistor from failure at the moment when the base of the transistor is present! negative half-wave of exciting voltage. In addition, in receive mode the cascade is closed and does not make any noise.

The choice of stabilization voltage is made from the condition:

U st< = U n k-e add.

Let's calculate the power of the modified cascade.

U" k-e = U k-e + U cm = 25 + 35 = 60B< U k-e add.+ 65V;

U to exc = U" to-e - U to-e us. = 60 - 5 = 55V;

I k a max = U exc x S = 55 x 46 = 2.53A;

P suspension = I a max x A o = 2.53x0.33 = 0.84A;

P supply = efficiency x P supply = 1000 W;

R a race = R lower - R n = 1680 - 1000 = 680< Р а доп = 700 Вт.

Thus, we see that compared to the original version, the power has almost tripled. In this case, the power reserve of the lamps is almost completely used.

It should be noted that this cascade operates with grid current. From which it follows that the grid voltage source must be stabilized and have sufficient load capacity - 200 mA. For some reason, the opinion is firmly rooted among radio amateurs that the grid current in the output stage lamp is almost a disaster. This is, of course, not true.

This opinion was probably established in those days when the vast majority of radio amateurs used lamps such as GU19, GU29, GU50, etc. Indeed, these lamps are not designed to work with grid current, since when the grid voltages enter the positive region, the linearity of the anode-grid characteristic is sharply disrupted. In addition, these lamps develop rated power even without grid currents. Another thing is metal-ceramic microwave lamps of the GI6B, GI7B, GS23B, GS35B, etc. series. These lamps are specially designed to work with grid current and develop rated power only when it is present.

In conclusion, a few words about measuring the output power of the “hybrid”. It will be limited only to controlling the anode current at the excitation peak, and then, taking into account the efficiency, calculating the output power in some cases will not always be correct. This is probably what the author of the mentioned article did.

The fact is that starting from a certain level of excitation voltage, the increase in the anode current continues, but the RF voltage at the load equivalent does not increase, often even drops. This is explained by the fact that positive half-waves introduce the transistor into a saturation state. This is not the appearance of a grid current, as can sometimes be heard on the air. For example, in the amplifier according to the circuit in Fig. 1, there cannot be a grid current in principle, but nevertheless this effect remains.

The greater the amplitude of the excitation voltage, the longer the transistor is in the saturation state, the resistance of the emitter-collector junction decreases more and more, the current through the lamp increases, but there is no increase in voltage at the equivalent. Therefore, in any case, the RF voltage at the equivalent load should be monitored. The cascade power should be set 10...15% below the maximum achievable, by correspondingly reducing the exciting voltage.

A few words about the design of the amplifier. There are no additional design requirements. The lamps are placed on a metal plate, which, in turn, is mounted on four high-voltage KVI capacitors with threaded fastening.

The capacitors are located at the four corners of the plate. Structurally, capacitors serve as support posts and, at the same time, are blocking ones. The input impedance of the output stage is approximately 30 Ohms. This circumstance, in a certain way, increases its stability, but requires the adoption of some measures in coordination with the previous stage of the transmitter or transceiver.

The parameters of the P-circuit, anode choke and other design features are not given, because the author focuses on the method of cascode connection of the amplifier stage.

G. Panov, (UA3AUP)

Literature:

1. S. Voskoboynikov “Power amplifier” - Radio amateur.

Hybrid audio amplifier, which is shown in the diagram below, is considered by many music lovers to be one of the best devices of this type, incorporating all the best that a tube and transistor UMZCH can provide to the maximum. Its sound is similar to a push-pull device made on triodes, but the bass is much richer, faster, clearer and more solid. The midband is transparent with pronounced details, the upper frequencies are free of any impurities that transistor devices suffer from. I've been thinking about building a high-end power amplifier for a long time. Having gone through various options for circuits, of which there are a great many on the Internet, it was this circuit diagram that attracted the most attention.

In general, as a basis, this schematic solution was absolutely suitable for me, however, later, as the setup progressed, the need arose to modernize it a little. The scheme is excellent, but there were not enough protective functions. Therefore, I first of all added protection that ensures a soft start of the amplifier when the mains voltage is turned on. Improved the function that performs automatic voltage bias on MOSFET IRFP140 and IRFP9140 transistors. In the original author's design, the voltage from the output of the lamps was significantly lost in a bias circuit with low internal resistance. Only after I increased its total resistance to several hundred kOhms did the output amplitude swing increase to 30v. p>

Ultimately hybrid amplifier provides output power of up to 200 W per channel when operating into a 4 ohm load. Based on the fact that the output stage of the device operates in class A, I pre-arranged the installation of heat sinks for field-effect transistors, and an additional fan to cool the radiators. In terms of technical and sound parameters, this circuit is very similar to the famous Magnat RV3 hybrid power amplifier. A significant difference between this amplifier and the Magnat is that the output stages of the latter use silicon bipolar transistors, while in this one the final stage operates on field-effect transistors. It was the use of MOSFET transistors that eliminated the need to install additional matching channels, using only capacitors as transition elements.

Speaking about devices of this type like tube-transistor amplifier, it is worth noting that the main goal is to obtain high output power, not for the sake of volume in the speakers, but to reproduce high-quality, natural sound. It is also worth noting another design feature of the device. To provide supply voltage to the tube amplifier module, a switching power supply was used with a constant output voltage of 6.3v and 270v, as a result of which it was possible to eliminate the low-frequency background as much as possible and radically reduce the noise level.

Important note! The diagram presented here, as stated above, was used as a basis. Therefore, everyone who may be planning to repeat it has the opportunity to improve it in their own way. I would also like to add that during the testing process I decided to completely remove the cascade installed between the capacitors and field-effect transistors. At the moment, a cascade is installed that sets the bias on the gates. The main elements of this cascade are variable, multi-turn resistors, as well as zener diodes; it may be necessary to replace the permanent stabilizers with adjustable ones.

Transcript

1 Circuit design of a hybrid amplifier. E. Vasilchenko, Kazan. June 2002 In this article, I decided to abandon the generally accepted rule of writing technical, scientific and pseudo-scientific articles that require presentation in the third person. Reflections on the role of sound-reproducing devices in our lives led me to the conclusion that the creative, emotional aspects of this problem are no less important than the technical ones (though not so much as to replace one with the other). In the world of technology, which is 100% formalized, there is no place for the author’s emotions. The scientific world has much more degrees of freedom; serious passions boil in it, and sometimes the academic lines “it has been studied, it has been shown” cause a storm of delight or indignation among initiates. This tradition, carried over into popular technical publications, played a cruel joke on low-frequency radio amateurs, largely predetermining the modern situation. While magazines in recent years have been talking about a vinyl and tube renaissance, it's time to wonder where we were all looking before? After all, there were people who never put signal transformers on the shelf and never threw lamps in the trash. I keep on my desktop, unknown how, a clipping from Radio magazine with an editorial article from 35 years ago with the subtitle “From the XI Scientific and Technical Conference at the IRPA” that came to me. Without comment, I will quote an excerpt: In the reports and speeches of the conference participants, the heads of individual enterprises, which still continue to produce receivers and radios, the cost of which is higher than the selling price, were sharply criticized. Radio industry enterprises face great challenges in the current five-year period. First of all, production volumes must be increased. If during the period 21.5 million radios and radiograms were sold, then in it is planned to sell 30 million. But the sharp increase in production volume and the task of selling products put forward demands for continuous improvement of models, increasing reliability and sound quality, improving their appearance, design, architectural forms, colors, ease of use, and reducing costs. This means that it is necessary to organize production in such a way, to find such technical and organizational solutions that would facilitate the rapid introduction into production of models that are in all respects at the level of world standards. The work carried out at IRPA and design bureaus of leading factories, as well as the experience of production activities of all enterprises in the industry show that these problems are solved through transistorization and unification of broadcasting equipment. During the period from 1966 to 1970, it is planned to convert all radios of the first, second and third classes to transistors. The only exception will be monophonic and stereophonic radios of the highest class, which will continue to be produced on tubes. Transistorization of household radio broadcasting equipment will significantly reduce its dimensions, increase reliability by 1.5-2 times and achieve significant savings in energy and materials. It is estimated that as a result of transistorization, savings due to reduced costs for materials per year will amount to 2.5–3 million rubles. In addition, 170 million kWh of electricity per year will be saved. Radio, 1966, 8, p. 21. “The focus is on transistorization and quality,” writes an author unknown to me. Every time I share my experience with readers or interlocutors, I remember this article. The creation of sound equipment is a unique area of human activity, where almost any person who knows how to handle a pin and a metalwork tool, to the best of his qualifications, can appreciate the value of the ideas embedded in the design. That is why the description or presentation of the plan must be personalized and separated from the opinion of the editors or fellow employees. The impersonal formula "one can conclude" must give way

2nd place to honest “I think”. To the best of my ability, I will try to implement the decisions of the mentioned conference with detailed comments. The history of the creation of the amplifier described here began quite a long time ago, more than 10 years ago. At that time, there was no domestic audiophile press, only a select few lucky people had access to the Internet, and libraries had already stopped receiving foreign magazines. The main and most popular source of information by inertia remained the magazines "Radio" and PTE (Instruments and Experimental Techniques). When almost all known transistor UMZCH circuits over the past 20 years have been repeated and tested by ear, the question arose: “What to do next?” It cannot be said that there was nothing worthy in the whole mass of schemes and designs. Each year brought a new leader. The first milestone in the mass transistorization of amateur designs was, undoubtedly, “High-quality amplifier”, S. Bat, V. Sereda. This was the first "people's" UMZCH. In essence, it was a high-power operational amplifier. The development of this topic now seems to me to be a dead-end branch. Not everything that is good for driving electric motors and other actuators is good for amplifying sound. This construction turned out to be unusually tenacious and was replicated in dozens of varieties, despite the poor sound. Transistor amplifiers of those years did not win the war with tubes. These lamps gave up key positions without a fight. Leafing through the “Radio” of tube times, one cannot help but be surprised at how well the authors implement the decisions of the mentioned conference. It’s just that high-quality tube amplifiers did not seem to exist, but “small-sized ULF”, “VLF with increased efficiency”, etc. were presented in abundance. The tube theme in mass publications was doomed, and a few years later young radio amateurs were perplexed when they encountered comparisons of this or that device with tube monsters. The deterioration of amateur transistor amplifiers of those years was no secret to anyone. But the developers worked tirelessly and at the end of the 70s there were already very decent-sounding amplifiers. Until 1965, most Telefunken, Grundig, Fisher amplifiers were made using tube circuitry: with interstage transformers, using germanium transistors of the same conductivity. After 1965, manufacturers gradually switched to silicon transistors. The characteristic circuit topology of that time is illustrated by Beomaster 3000, Uher CV-140. With the advent of powerful complementary transistors in the 70s, amplifiers began to be built using symmetrical circuits. One of the first representatives of this trend was the JBL amplifier, released in 1967. Subsequently, such circuitry was used by SAE, McIntosh, Hafler. At the same time, circuits with differential amplifiers appeared. It is curious that experts note the better sound of European amplifiers, which did not use differential drive of the output stage, unlike amplifiers from Japanese and American companies. By the mid-70s, integrated circuits (Braun A301) began to be widely used. The amplifiers mentioned deserve detailed analysis and even repetition. However, let's return to the schemes that domestic amateur designers could see and repeat. This is Quad-405, the diagram of which was published in Wireless World in 1978 and is familiar to us from O. Reshetnikov’s article in the December 1979 issue of Radio. Without a doubt, the most famous amplifier is Michael Wiederhold, first described in 1977 in Radio fernsehen electronik. This scheme is still published in various variations ("Radio" 4/78, "Radio" 6/89, "Radio" 11/99). Thanks to the work of M. Otal and Marshall Leach over the years, amplifiers have gotten rid of one type of specific TIM distortion caused by the limited speed of transistor stages. Around the same time, works by A. Mayorov and P. Zuev appeared on dynamic distortions in amplifiers. Many people remember A. Vitushkin’s good, although not the simplest, amplifier from the July 1980 issue of Radio. A. Syritso's bridge amplifiers sounded very good (especially from Radio 11/82). Many interesting circuit solutions were published in PTE collections. As high-voltage and high-frequency pnp transistors appeared, amplifiers became increasingly broadband, powerful, and linear. However, the problem of unsatisfactory sound in general remained

3 solved. The article “The phenomenon of transistor sound” added fuel to the fire. Let me remind you that the authors compared various amplifiers in a fairly good environment (studio sound reproducing devices and a professional spectrometer) and, based on their observations, made conclusions that are presented here: From all that has been said, the following conclusions can be drawn: - “transistor” sound is not a mandatory property of transistor amplifiers LF; its nature seems to lie in the imperfection of these amplifiers; - “transistor” sound disappears when the harmonic distortion decreases to 0.03–0.04% over the entire operating frequency range; - with a modern element base, the specified limit of the harmonic coefficient is achievable only with a sufficient depth of overall environmental protection. Now, when the number of our own amplifier developments has exceeded the second ten, it is easy to blame the authors for the incorrectness of the problem statement, but 20 years ago, it seemed to me, like so many amateurs, that a recipe for good sound had been found. You can simply, without paying attention to the long tail of distortions, suppress them with deep OOS plus some additional measures. The "race of zeros" began. The eighties became a dark period for sound circuitry. In order not to be unfounded, I will comment on the above quotes. The authors were looking for the “transistor sound phenomenon” in amplifiers with deep feedback, and this is akin to searching for a black cat in a dark room. It seems that if the kit had additionally included one amplifier, tube and transistor, without a common OOS, the results of the experiment would not be so unambiguous. The first comparison leader was the amplifier assembled by M. Leach. This is not surprising, it is truly the best in this class (that is, in the class of high-power operational amplifiers). In addition, M. Leach himself especially noted the role of the amplifier’s power supply, or more precisely, its ability to provide high current. Nobody took into account this most important feature of his amplifier. And a few more points that few people paid attention to at that time. Such a sound characteristic as “transistority” is subjective, and it is simply incorrect to extend the experience of one’s own perception to all listeners. And most importantly, the absence of a feeling of “hardness” or “transistority” is necessary, but not at all sufficient for a high-class amplifier. Readers of the modern audio press can easily name a dozen more signs by which sound quality is assessed. Rice. 1. Amplifier by Yu. Mitrofanova. The amplifier of Y. Mitrofanov, the circuit of which is given in the article, according to the authors, sounds better than all the others. It's not difficult to explain. The voltage amplifier (VA) of this UMZCH, Fig. 1, performed on V5, V6 has a small own SOI (0.15%) and a fairly large

4 power. The parallel OOS circuit has the minimum possible length; it is much shorter than in traditional amplifiers and is fed to the inverting input of the UN. The intrinsic nonlinearity of the output stage is also relatively small. This output stage was used in the famous QUAD 303 and in Brig. If we add to this a powerful low-impedance power supply, then these factors are quite enough to make the amplifier sound. And the SOI value of 0.02% is only a consequence of the topology features, and not the reason for good sound. Thus, the conclusions of the authors of the articles are, as in a joke about a mathematician, as accurate as they are useless. The race of zeros reached its peak in 1989 with the publication by N. Sukhov of the famous high-fidelity UMZCH, the basis of which is the M. Wiederhold amplifier, Fig. 2. Fig. 2. M. Wiederhold amplifier, 25-watt version. It has been repeated and continues to be repeated (using a modern element base) by thousands of amateurs. The range of reviews about its quality is very wide, and this is natural. How many people have so many opinions? Everyone's kits are different. Most are very pleased with them. Many claim that they have never heard anything better. I'm sure this is true, but what about those who are dissatisfied? And there are many of them; These are, first of all, owners of tube equipment, good acoustic systems, and simply experienced listeners. Let's try to figure out what's going on here. Undoubtedly, everyone’s capabilities are different, audio salons are not available everywhere, and modern “branded” equipment is not at all a role model (rather, on the contrary). The first thing that comes to mind is that all listeners have different speaker systems. N.E. Sukhov himself considers it his merit not so much to create the amplifier circuit as to equip it with a device for compensating wire resistance. It is possible that the influence of wire resistance on the damping of the AC cable PA system is relevant for amplifiers with zero output impedance, but not all amplifiers have output impedance formed by negative feedback. In addition, it would be a mistake to assume that the sound character of a complex is determined only by the damping coefficient. The main complaints of “listeners” about the sound of UMZCH VV relate to the accuracy of transmission of medium and high frequencies, where electrical damping of the speakers by the output impedance of the amplifier does not work. It is often said that this compensator “smears” the sound. At medium and high frequencies, nonlinear effects arise in speakers, which no device for generating output impedance can cope with. S. Ageev wrote about this in detail in. Differences in the design, packaging, and parameters of the power supplies of this amplifier also do not allow a correct assessment of the UMZCH VV through a “popular vote.” Those who want to get an idea about it can only resort to the most reliable way to listen to it themselves. This was done. The most promising amplifier of the 80s was

5 is assembled in metal in compliance with all rules for installing such devices. Comparison with other homemade products did not reveal any advantages of the UMZCH VV. Amplifiers by A. Syritso (N11/82) and according to the circuit from E. Gumeli’s article (N9/85), made by me several years earlier, sounded much more natural (and at the same time different in the midrange) with the same configuration and design. By the way, the measured SOI of all these amplifiers did not exceed 0.02%. Confidence in the correctness of the chosen path was shaken. New ideas were needed. First of all, it was decided to check the influence of OOS and various circuit topologies. Deep negative feedback was the first to be blacklisted. The prototype of the amplifier with number 6 was the “Amplifier without general feedback”. The authors used well-known components in circuit design: a push-pull emitter follower at the input, a push-pull scaling current mirror as a voltage amplifier, and a composite emitter follower at the output as a current amplifier. They very elegantly bypassed the problem of DC drift at the output of the amplifier, placed a large-capacity electrolytic capacitor at the output and used a single-polar supply. Perhaps, if I had had Black Gate or Elna Cerafine audio series capacitors then, this solution would have satisfied me. The best “electrolytes” then were K50-18 and I didn’t want to use them at all. Getting around this problem proved difficult. The amplifier was switched to bipolar power supply, the output capacitor was eliminated. To obtain more power, the voltage was increased to 2*30 Volts, the element values were recalculated, and the bias circuit was replaced with a traditional one (Fig. 3). Rice. 3. Amplifier without general feedback (6) Along the way, it turned out that the amplifier works better (more stable) with conventional rather than composite transistors. The struggle with the zero offset at the output began. A voltage amplifier assembled using a current mirror circuit is very sensitive to all disturbing factors: instability of power supplies, temperature and its gradient inside the structure, variation in the values of the elements and, most importantly, to the parameters of the transistors. If we calculate the overall gain of the voltage amplifier using the approximate formulas given in the article, it will be equal to approximately 7 (for an output power of 25 Watts). In fact, with this coefficient, the power ripple or, in the case of bipolar power supply, the difference in ripple of the positive and negative poles is transmitted to the output (not counting the useful signal, of course). It is for this reason that the authors of the circuit used the R19C5 filter in the power supply. Let's consider the cascade on VT4 (6). Its gain is approximately equal to the ratio of resistors in the collector and emitter, that is

6 R15 K u = 100. Therefore, the slightest drift in the emitter-base voltage of any of the transistors R 12 included in the cascade will lead to a significant change in mode. If this drift is caused by a general change in temperature in the case and the temperature of all transistors changes simultaneously, then the change in current VT4 and VT6 will be the same in magnitude, opposite in direction and will not lead to a change in the output potential. This is only possible in the ideal case, when transistors VT4 and VT6 are completely identical. In practice, there are no two identical transistors, much less with different conductivities. The difference in the values of h 21 E and U BE of the transistors of the cascade will lead to a significant difference in the collector currents, and, consequently, to a zero shift at the output. If you use the transistors recommended in the article without selection, then most likely the bias will be about 0.5 1 Volt at best. Moreover, when the temperature inside the case changes, the bias will also change due to different temperature drifts of the transistor parameters. In addition, the gain and output voltage of the AC arms will also be different. To some extent, this difference in gain can be compensated for by trimming resistor R9. It is impossible to balance the voltage booster for direct current by changing the resistors included in the cascade, since this will change the gain for alternating current. The cascade load consists of two parallel-connected branches, linear and nonlinear. Resistors R15 and R17 form a linear low-resistance (about 5 kohm) branch. The gain, efficiency and output impedance of the cascade are determined by them. The input impedance of the final stage is very nonlinear, but much higher (at least 100 kohm). Therefore, the component of the voltage output current that goes into the nonlinear load branch is relatively small, a few percent, and can be ignored. Let us examine in more detail the operation of the voltage amplifier stage. The DC operating mode is set by the resistance value R10. The current through it U is approximately equal to 1.2 mA: I R 10 =. The properties of the scaling current mirror are such that R10 IVT 3 R12 = = 3. Therefore, the current through transistors VT4 and VT6 is 3.6 mA. The magnitude of the quiescent current IVT 4 R 11 must be selected in such a way that when the current through the transistor changes under the influence of a signal, its gain remains, if possible, unchanged. The dependence of h 21 Oe on the emitter current is one of the two main reasons for the occurrence of nonlinear distortions in transistor cascades. Therefore, when choosing transistors and their operating mode, the corresponding characteristics should be taken into account. Unfortunately, then, more than 10 years ago, documentation for transistors was practically inaccessible to amateurs. Therefore, the mode had to be selected approximately to minimize distortion at the output of the entire amplifier. The maximum output voltage of the cascade is close to the supply voltage. Consequently, the alternating voltage at the output of the UN can be about 20 volts in our circuit. In practice, after 15 volts, soft limitation already began. This is due to the insufficient quiescent current of VT4(6), but it was fully consistent with the power of the speaker systems of 50 Watts. By increasing the quiescent current to 5 or even 10 mA, the power and linearity of the amplifier should increase, but such a goal was not set. The gain of the cascade on VT4 is about 100, which means 0.15 Volts are applied to the base of VT4. Let's check: 15 V at load R15 = 10 com will be at a current of 1.5 mA. This means that the alternating current VT4 is 1.5 mA, and the signal voltage drop across R12 = 100 Ohm will be 0.15 V. To find out what part of this voltage is applied directly to the base-emitter junction, remember that the volume resistance of the emitter of the transistor is directly proportional to the temperature and inversely ϕt to the current: rе =, where ϕt is the so-called temperature potential, at room temperature IE is approximately equal to 26 mV. With a constant current through VT4 equal to 3.6 mA, its emitter resistance will be 7 Ohms. An alternating current of 1.5 mA will create a voltage drop across it

7 10 mv. Another useful relationship is that each millivolt of AC voltage applied to a pn junction adds 1% of the second harmonic level to the output current. With such a signal at junction VT4, the output current will contain 10% distortion. Local negative feedback is created through 100 Ohm resistor R12. Its depth is equal to the ratio of resistances R12 and r E, that is, 100/7=14. This OOS reduces the level of the second harmonic by 14 times. That is, transistor VT4 in this mode introduces 0.6% distortion. In push-pull cascades, even harmonics must be compensated, provided that the cascade is completely symmetrical. In reality, shoulder reinforcement always varies slightly. Therefore, we can assume that the level of the second harmonic is from zero to 0.3%, depending on the degree of symmetry. The level of the third harmonic with such a signal value at the transition is usually several times less than the level of the second and it is not compensated. You can expect its level to be 0.03-0.06%. At high frequencies, the asymmetry of the arms increases and compensation of even high-order harmonics is not as effective. The second source of distortion is the nonlinearity of the base current VT4. It can also be estimated from a graph of gain versus current. Since we do not have the required data, and the domestic industry is not very kind to developers, we will use typical values for imported general-purpose transistors. For example, take a pnp transistor 2N3906 from ROHM. In terms of parameters, it is approximately equivalent to (or better than) KT3108 and KT313. According to graphs from the company's website, when the emitter current changes from 1 to 4 mA (that is, by 300%), h 21 Oe changes from 110 to 140 (by 25%), Fig. 5. This is a significant nonlinearity; modern transistors for audio applications have much better characteristics. Rice. 5. Dependence of the gain of the 2N3906 transistor on the collector current. Typical for small-signal cascades, the change in the emitter current is % of the quiescent current. In other words, during the signal period the base current transfer coefficient changes by 0.5 1%. The base current also changes accordingly. In our case, the base current is I E 3.6 I B = = = 30 µA. The nonlinear component of the base current, equal to 1%, will be 0.3 μA. h21e 120 The alternating current of the VT4 base, flowing through the output resistance of the previous stage, creates a voltage drop across it applied to the base, and this voltage contains a nonlinear component. The output impedance of the previous stage is determined mainly by the R8R9 circuit. The output resistance of the composite emitter follower VT1VT2 is a few to tens of ohms and can be ignored. The nonlinear component of the base current VT4 flowing through circuit R8R9 will create a voltage drop on it of 0.3 µA * 3.3 kom = 1 mV. This is the amplitude value, peak to peak. The effective value is less by 2 2, or approximately 3 times, i.e. 0.3 mv. As we remember, the useful signal based on VT4 is 150 mV, therefore, the base current already contains 0.3/150 = 0.2% distortion. Everything that was said about compensating for even-order distortions also applies to base currents.

8 A quick analysis of the operation of this voltage amplifier gives us the opportunity to draw some conclusions. The first and obvious one: in the author’s (magazine) version of the amplifier, the transistors operate in a non-optimal mode. To increase linearity, the quiescent current of the cascade should be increased several times, because even at 10 mA the dissipated power will not exceed the maximum permissible. The second conclusion concerns the choice of transistors for such a circuit. These must be modern high-linear transistors. KT313 and KT3117, and even more so KT502/KT503, are not complementary pairs. With them it is almost impossible to obtain an acceptable SOI. Complementary pairs must be carefully selected according to h 21 E and U BE. Only in this case can the stability of the operating point and a low level of distortion be ensured. Additionally, the thermal stability of the operating point of the voltage amplifier can be ensured by design measures. The printed circuit board had to be positioned so that all four transistors were nearby and could be covered with a cap. Without it, any breeze blowing on the board would cause the output zero to drift. I managed to increase the potential at the output of the amplifier channels to 25 and 50 mV without using additional balancing. The third conclusion may seem somewhat unexpected, but we should not forget that this small study was launched with the aim of understanding the influence of OOS on sound. In my opinion, introducing general OOS into such an amplifier not only makes no sense, but is also harmful from the point of view of sound quality. Feedback can cover cascades that are initially linear, and then it will fulfill its purpose. Namely: it will ensure the stability of the circuit parameters over time and under different operating conditions. In the analyzed circuit, this stability is ensured parametrically, that is, by using components with precisely specified parameters. If the component parameters are chosen at random, the circuit will become unbalanced and become a source of distortion. Using OOS to correct this curvature only leads to a change in the spectral composition of distortions towards increasing the number of harmonics, but not to their elimination. The higher the degree of symmetry of the original amplifier, the less “work” there will be for the OOS. To realize all the capabilities of this voltage amplifier, I had to re-wire the printed circuit board several times and change the design of the amplifier. In intermediate versions, the UN was even placed in a thermostat. The most difficult thing was to select four pairs of complementary transistors. After futile attempts to select such pairs from KT3117, KT313, KT3108, KT502, KT503 using a simple stand and tester, I took 50 pieces of unknown Korean transistors S8050, S8550, also known as S8050, S8550. It was not possible to find their characteristics, so I looked into the incoming control department of one of the factories. Armed with an automatic transistor tester, I checked the maximum permissible voltage between the collector and emitter and sorted them by h 21 E and U BE. An increase in the reverse collector current began at voltages above 110 V. The base current transfer coefficient turned out to be within the limits for both n p n and p n p transistors. When the emitter current changed within 1 10 mA, h 21 Oe changed slightly. After that, selecting pairs and finishing the amplifier turned out to be a very simple matter. I did not specifically configure the output emitter follower with a shunt compensator, limiting myself to selecting the quiescent current of the output transistors to minimize distortion. At a current of 300 mA, the automatic nonlinear distortion meter S6-11 showed a minimum of about 0.1–0.15%. Each amplifier channel was powered by a parametric stabilizer, Fig. 6. The heating of the stabilizer transistors is insignificant, so it turned out to be possible to attach the corners on which they are installed directly to the duralumin bottom, through a mica gasket. The amplifier's printed circuit boards, measuring 70 x 80, are screwed directly onto the heatsinks of the output transistors, which have an area of 600 square meters. see the channel. The radiators have good thermal contact with the bottom and massive front panel. The heating of the amplifier during operation does not exceed 60 70

9 degrees. 80 Watt toroidal power transformers are separate for each channel. Rice. 6. Amplifier power supply 6. Listening to the amplifier showed that the time spent searching was not in vain. The amplifier had an extremely soft and delicate voice. The mid-frequency range was especially good. The resolution and sound detail were higher than all of its predecessors. He softened the highest registers, while the traditional ones, “from the Radio,” simply turned them into “sand.” Despite the packaging, which is completely unsuitable by today’s standards (K73-17, K50-18 and not the best transistors), this amplifier still has no competitors in sound quality among the so-called “affordable high-end” and delights its owner with the opportunity to listen to his favorite recordings , not test discs. It must be admitted that the experiment turned out to be very informative. The experience gained in the design of amplifier 6 without general feedback set the direction for further developments. The analysis of the circuits and the listening results are quite consistent with the modern unspoken rules of audio circuitry. In recent years, when the Internet has turned from a symbol of an incomprehensible luxury into a necessary tool, do-it-yourselfers have had an excellent opportunity to communicate and exchange experiences, both among themselves and with professional developers. The specifics of using transistors in sound circuits are gradually becoming available to a wide range of DIYers. There has never been a single recipe for building a good amplifier, but there are some general principles that most designers come to sooner or later. All developers assess the importance of a particular principle differently; This scale of values is not linear, constant and absolute, because it depends on many subjective factors. Therefore, I present my own list, based on more than 20 years of experience in building amplifiers, of the most important requirements for the design of UMZCH in descending order of importance. Of course, no one is stopping the designer from sacrificing any of the items on the list to some additional idea. A) The power source must provide the final amplifier with a current that is both powerful and clean. In modern interpretations, the amplifier is often represented as a current modulator. Therefore, the quality of the current supplying the output stages must be as high as the development budget allows. The power supply is a full-fledged participant in the audio path with all the ensuing consequences. Any secondary power source contains reactive elements that form filters. For filters, parameters such as transient response, quality factor, and characteristic impedance are defined. The influence of these factors on sound is practically not considered in the literature. But these are well-known, easily measured and at the same time parameters that greatly influence the sound. B) One of the most important components is the voltage amplifier. Perhaps this point is not as obvious as the previous one, and not all amplifiers are built according to the UN UT circuit, but many designers note that both tube and transistor output stages are “transparent”

10 for sound, and the “voice” of the amplifier is determined by the driver stage or NA, respectively. Human hearing, especially trained hearing, has an extremely high sensitivity to the spectral composition of distortions. Small differences in the power of even and odd harmonics, differences in the rate of decrease in spectral density, and the presence or absence of dominant harmonics are perceived as a change in the character of the sound. In an ultrasonic amplifier, the dynamic range of the amplifier element is usually fully used and the operating point covers the largest portion of the amplitude characteristic. Its nonlinearity manifests itself most clearly here. Therefore, all elements have their own spectrum of distortion, a kind of barcode by which they are unmistakably recognized by ear. C) The number of cascades should be minimal. It doesn’t matter whether it’s transistor or tube, but each additional stage introduces additional nonlinearity. There are many reservations in this point, as well as in all the others. Getting maximum gain from a stage can degrade stability and with it linearity. This means that there is a certain balance between the depth of the local OOS and the magnitude of the cascade gain. The designer's task is to find a compromise. D) The quality of components, both active and passive, must be adequate. An absolutely indisputable point. The only question is what is considered important and what is secondary. Most often, this question is closely related to the degree of hearing training and the thickness of the wallet. D) Thoughtful design and temperature conditions. We are talking primarily about vibration isolation, since most radio elements have a noticeable microphone effect. Calculating sound fields in devices is very complex, so designers usually use empirical data and their own experience. The temperature inside the case not only affects the service life of the elements, but also significantly affects the sound. The formation of these principles for me began precisely with the experiments described above. In the next development, I decided to test the effect of the principle of minimalism on amplifier 8 (number 7 was a tube amplifier corrector for a vinyl player). The assembled UMZCH VV boards remained from previous work, and they became prototypes for studying the nonlinearity of various cascades. The first test subject was the output emitter follower; it then entered the circuit of the proposed amplifier without changes, Fig. 7. Fig. 7. Amplifier output stage 8.

11 Circuit design analysis. The quiescent current of all three stages is set by resistors R3, R4, and is regulated by a variable resistor R2. The VT7 transistor is traditionally mounted on the heatsink of the output transistors and performs the function of setting and thermally stabilizing the quiescent current. Resistors R6, R7 are added to ensure stability of the amplifier during tuning when the length of the connecting wires is long enough. Sometimes the same resistors are required in the bases of the output transistors. Typically, the output stage is connected to the voltage amplifier either by the upper (according to the circuit) or the lower shoulder. The first stage of the repeater always operates without cutoff, in class A. The same signal current flows through VT1 and VT2, the voltages at their emitters must be exactly equal in amplitude. Therefore, it is considered acceptable to excite the output stage in one arm. This is correct only for traditional circuit design - when the transistor that sets the bias (VT7) is located in the collector circuit of the voltage amplification stage. The UN has a large output resistance, especially when connected with a common base, and is usually (if the circuit is asymmetrical, that is, excited only on one side) loaded onto a current source that has an even larger output (megohms). Therefore, there is practically no current through transistor VT7. We had to replace current sources with resistors. Under these conditions, a noticeable alternating current flows through the stabilizing transistor VT7. Therefore, its dynamic resistance and its nonlinearity can no longer be neglected. The direct current through this transistor is approximately equal to 1 mA (the current-setting resistors are 43 kΩ from the 44 V supply). The transistor itself is turned on with a gain of 6 times, since it sets the bias to 6 p-n junctions. Therefore, its dynamic resistance in such a connection is 6 times greater than the resistance of its emitter. As already mentioned, at this current the emitter resistance is 25 ohms. We find that the AC resistance of VT7 is 150 Ohms. This means that the signal is supplied to the second arm slightly weakened, by 3.5% (150 Ohm/43 kOhm = 0.035). This gives about 0.17% even harmonics. Capacitor C2 is turned on to bypass the dynamic resistance VT7, and this significantly reduces the THD. It would be more correct to send a signal to both shoulders at the same time. In conventional amplifiers (i.e. DC op amps), shunting also improves performance, but this comes at the cost of improving the symmetry of the RF base circuits. Blocking the phase difference in the halves of the push-pull stage suppresses distortion caused by unequal delays in the arms. When the output stage is powered at 44 V, the maximum amplitude value of the output signal will be approximately 4 volts less. This drop is the sum of the saturation voltage of the output transistors (about 1 1.5 V), the drop across the emitter resistors R9, R10 (also about 1 V). In addition, 0.65 V will remain at the emitter junctions of all three stages: after all, the signal voltage based on VT1 should not be higher than the supply voltage in order to avoid breakdown of the collector junction. The amplitude value of the output voltage of 40 V into a 4 Ohm active load will give 10 A collector current. This is a lot for the selected type of transistors. At this current, the cutoff frequency and gain of the transistors drops significantly. Transistors remain relatively linear up to a current of 2–3 A. Even the best imported transistors, specially designed for audio applications, lose their amplification and frequency properties when the collector current increases above 5–6 A. In addition, when the collector-emitter voltage decreases to several volts, the capacitance of the collector junction increases ten times or more. Therefore, it is undesirable to use this stage in this mode due to high distortion. The output power 2 U m will be P = 200 W, if the power supply allows. Each transistor in this 2 Rn 2 1 U pit case dissipates Pdiss = 50 W (in class B), which is quite acceptable if there are 2 π Rn sufficiently efficient radiators. But still, the amplifier works much better at an 8-ohm load, this is confirmed by measurements. If the load has a reactive component, then the dissipated power and collector currents increase.

12 The base current transfer coefficient of high-quality output transistors is usually in the linear region and up to in the high current region. For domestic transistors these values are slightly lower, 1.5 2 times. For calculation purposes, the minimum values are usually taken, because in the production of equipment, the selection of components is usually not allowed. Nobody will stop us from selecting transistors based on their gain and setting standard, not minimum, values. Despite the fact that the transistors in the emitter follower are covered by 100% negative feedback, it is better to ensure symmetry by design measures. The amplitude of the base current will be Ib = Ie / h21e = 10A/30 = 0.3 A. The pre-output transistors must supply this current. In real operating conditions, the current amplitude of transistors VT3, VT4 does not exceed 100 mA, but this is also a lot. At this current, few medium-power transistors can operate in the linear portion of the characteristic. Among the domestic transistors there are no such ones that would have an extended section with a constant h 21 Oe, have good frequency properties and would be complementary. Therefore, it is necessary to use either very low-frequency and nonlinear KT850/KT851, or, when power is reduced, KT940/KT9115 or KT639/KT961. Both of them are not complementary pairs, since they have significant differences in gain factors and cutoff frequency. Looking ahead, I note that transistors for output stages of TV or computer displays have good frequency properties and high linearity, such as 2SA1380/2SC3502 from Sanyo. They will be very good in the first emitter follower stage. If this amplifier were being made now, I would put available imported pairs 2SC1837/2SC4793 or 2SB649/2SD669 into the second stage. The output could have been Samsung TIP41C/TIP42C, Toshiba 2SA1302/2SC3281, Mospec or SanKen 2SC2922/2SA1216, Motorola MJ15003/Mj15004, etc., but at that time they were not available. In addition, I was interested in the contribution of each component, so the transistors were not selected according to parameters, only those with low gain or noticeable leakage were rejected. Power was supplied from an unstabilized source of sufficient power. The first question that had to be resolved was what quiescent current to set. To do this, a signal from the G3-118 generator, which has fairly low intrinsic distortion even without additional filters, was supplied to the input of the emitter follower. The amplifier was loaded with a resistive load equivalent of 4 or 8 ohms, and the signal was monitored by an oscilloscope and an automatic nonlinear distortion meter S6-11. Most measurements were made at a frequency of 1 kHz. At a quiescent current of 100 mA, the current amplifier showed a stable SOI result of about 3% over almost the entire power range. And only for a small signal, when the output transistors operate without cutoff, in class A, the harmonic distortion drops to 0.5-0.6%. By increasing the quiescent current to 3 A, we get 0.6–0.7% of the output power up to W. It’s worth making a big digression here regarding crossover distortion. On a small signal, while the signal current through the transistors (or lamps) is less than the quiescent current, the transistors of the arms work on the load simultaneously, then when the level increases, one of the transistors closes. This is equivalent to doubling the output impedance. That is, the dynamic characteristic has a sharp break. You can “see” crossover distortion this way: by connecting and disconnecting the load at a low level, use an oscilloscope to detect the “drawdown” of the output signal. Then increase the level and do the same operation. While the reinforcing elements work simultaneously, they practically do not notice the load change; when moving to class B, the drawdown is more noticeable. In practice, the mechanism is somewhat more complicated, since the output resistance of the transistors depends on the current through them, in addition, stabilizing resistors R9, R10 are connected in series with them. The value of these resistors greatly affects the amount of crossover distortion. There is some resistance, which at a given quiescent current provides a minimum of distortion. The optimum is obtained when the output impedance of the entire amplifier changes least during the transition from a small signal, when both arms are active, to a large one, when one arm is closed. That is, it is necessary to calculate the output resistance for a small signal (output voltage is near zero) and for a large one, when the emitter current is greater than the current

13 rest several times. For powerful transistors, the simplified formula for calculating the resistance of the emitter body is not applicable; domestic transistors have never been accompanied by such data, so we will use data from the Internet. The website of the Danish company LCAudio provides a description of the amplifier The End Millenium. This is an amplifier without general feedback, so everything said above also applies to it. The output stage uses 200 watt SanKen 2SC2922 and 2SA1216, one of the best modern output transistors. I will give a table of the dependence of the emitter resistance on the load current, taken from there. The main feature that distinguishes these transistors is the relatively slow decay of output resistance at high currents, which is very useful for reducing distortion. Other high-power transistors have much lower output resistance (as well as gain and cutoff frequency) at high currents. Table 1. Load current Resistance, Ohm 100 ma 0.2 500 ma 0.10 1A 0.09 5A 0.08 10A 0.07 At a small signal, the output resistance of the amplifier will be m 1 1 Rout = (Rtr + R9) = (0 .2 + 0.1) = 0.15 Ohm, 2 2 B On a large signal R = R + R9 = 0.09 + 0.1 = 0.19. The difference, although not twofold, is there. out tr Consequently, there are also nonlinear distortions caused by a break in the dynamic characteristic. Let's calculate other combinations of quiescent current and resistance of stabilizing resistors. The linearity criterion will be the relative increase in output resistance during the time the current increases from zero to maximum: drout = (rb-rm)/rm in percent; The transistor resistance is obtained by interpolating the tabular data: Table 2. Current, ma R9, R10 Rm, Ohm Rb, Ohm drout, % ,1 0.15 0.17 0.1 0.12 0.17 0.2 0.17 0, 27 0.1 0.1 0.17 0.1 0.17 0.18 0.1 0.1 0, As can be seen from the table, stabilizing resistors greatly influence the nonlinearity of the output resistance. Their influence is greater, the higher the quiescent current is selected. The output resistance of the amplifier changes the least without these resistors (line 6) and The End Millenium (line 1). In the article “Current dumping: does it really work?” (Wireless World, 1978) Vanderkooy and Lipshits especially emphasized the advantage of class B amplifiers - they have no crossover distortion. I think that a simple Current dumping amplifier (Radio N9, 1985), like the famous Quad 405, is not bad sounds exactly why. Concluding the analysis of this part of the circuit, I note that “seamless” joining of half-waves is possible if the transistors have ideal (that is, logarithmic) current-voltage characteristics, and the emitter and base resistances are equal to zero. If the voltage at the base junction of one of the transistors increases by 100 mV, the emitter current will increase 10 times. In this case, the voltage at the second junction

14 transistor will decrease by 100 mV and its emitter current will decrease by 10 times, but will not stop. The overall characteristic is not linear, but there is no sharp break leading to the appearance of high-order harmonics. In real conditions, the resistance in the circuits of the transistor electrodes has a non-zero value, so the decrease in the emitter current of the closed arm occurs faster than according to the logarithmic law. Therefore, the switching of the arms occurs faster and, most importantly, with a complete cutoff of the current of the arm being closed. If no additional measures are taken, switching distortions are of a high order and are practically not attenuated by the OOS circuit. The consequence of all that has been said is the presence of a certain region of the optimal regime. This is intuitively guessed without any thought experiments. However, most often amateurs make the wrong conclusion, believing that the quiescent current should be as high as possible. In fact, the optimal quiescent current of the output stage depends on many factors, among which the decisive ones are the resistance of the emitter resistors and the parameters of the transistors used. Of course, if the entire amplifier operates in gain class A (that is, the current through the transistors never stops), many of the problems described are automatically eliminated. But still, true class A is quite difficult to implement in high-power transistor amplifiers. One problem is replaced by another. An indirect indicator of complexity can be the almost complete absence of such amplifiers on the market. The only monsters that come to mind are Mark Levinson, AM audio, Accuphase A50, Nelson Pass single-ended amplifiers, and the old 12-watt Sugden A21. Many manufacturers, declaring amplifiers as “Pure class A”: Plinius SA100, SA102, SA250, Musical Fidelity A2, etc., are clearly wishful thinking. Just look at the dimensions, weight, radiator area and power consumption to be convinced of this. Most likely, they operate in class A up to watts of power, like the top models from Pioneer, Sony, etc. The problem of thermal stabilization and energy supply of the cut-off mode at output powers W is solved quite simply. When trying to obtain more power, the designer is faced with the task of ensuring the normal operation of all components over the entire temperature range of operation, as well as with a sharp increase in the cost of the entire structure. Therefore, the vast majority of industrial amplifiers operating with high quiescent current have a break in the amplitude characteristic in the region of medium powers. As has already been shown, the higher the quiescent current, the more the output resistance changes when switching. This change is a prerequisite for the occurrence of distortions. All efforts of designers are aimed at optimizing the switching speed of transistors. In this case, the spectrum of distortions moves to the low-frequency region, where they are quite effectively suppressed by the negative feedback. An abundance of trademarks “class A+”, “AAA”, “economical A”, etc. indicates the marketing attractiveness of the “Class A” badge, but even the simplest calculations indicate that the least problems will occur with a reasonable choice of the quiescent current at the ma level. Let's return to our diagram; the smallest integrated SOI of the final amplifier was obtained at a quiescent current of ma. Without a weighing filter it is about 0.5%. Most likely, by selecting the value of the emitter resistors and the quiescent current, this value can be further reduced. The pre-output cascade operates with a quiescent current of 35 mA. Signal cutoff in one of the arms occurs when signal currents are close to the maximum, that is, most of the time the cascade operates in class A. Of course, switching the transistors of the pre-output stage also changes the output current and causes distortion. Typically, designers try to transfer the switching moment to the region of statistically rare amplitudes. The first stage of the current amplifier has a quiescent current of 4 mA. This is enough to ensure that the current through the transistors is not interrupted over the entire range of signals and loads, including during a short circuit of the load. The mode of this stage is selected as usual, in the region of a stable gain of the applied transistors. Before moving on to the analysis of the input stage, I will note the role of the Bouchereau chain R11C3. Its task is to ensure a favorable load of the output stage at frequencies above audio frequencies, that is, more than 50 kHz. At HF, the load (speaker systems with cable) always has a reactive nature with random

15 module and phase. Therefore, various RLC circuits are used to match the amplifier and the RF load. The best results are provided by a two-link chain like . As already mentioned, the composite emitter follower VT1-VT7 has a sensitivity of about 35 V rms. Its input resistance is almost completely determined by resistors R3, R4, connected in parallel with alternating current. Thus, the input resistance does not depend on the signal amplitude (which has a beneficial effect on the linearity of the amplifier) and is com depending on the value of R3, R4. Power consumed by the final U input stages from the voltage amplifier: Pc = = 0.06 W. Rin 20k The choice of an ultrasonic electronic tube as an amplification element is justified mainly by the simplicity of the solution and the predictability of the result. It would be possible to use semiconductors, but, firstly, this was already tested in previous work, and secondly, the microcircuit-transistor UN, with which this output stage previously worked, did not prove itself to be the best. To check the linearity of the voltage amplifier, we will assemble a rheostat cascade on a triode with a common cathode, Fig. 8. Fig. 8. Rheostatic triode stage. A signal from a sinusoidal generator with a voltage of 1–3 V is supplied to the input of the cascade. Resistor R4 is a load resistor. The voltage from it is supplied to the nonlinear distortion meter. The purpose of the experiment is to select a lamp that allows you to obtain the highest output voltage with minimal distortion. The anode resistance to alternating current in this circuit is less than 7 kohms, so the internal resistance of the lamp must be much less than this value, otherwise it will not be possible to obtain sufficient gain. To study the cascade, the input voltage is gradually increased until the level of nonlinear distortion begins to sharply increase. The peak output voltage (as measured by an oscilloscope) and the SOI level are recorded. Table 3. Lamp Quiescent current, mA Uout.max. (Peak)V SOI, % 6N6P N23P N1P Table 3 shows the measurement results with some widely used lamps. As you would expect, low-resistance ones allow you to get higher voltage. Therefore, 6N23P was chosen, which also has a relatively high gain. Despite

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New type hybrid HF power amplifier

Radio amateurs using professional radio receivers experience difficulties in obtaining in the transmission path the power of several tens or hundreds of watts necessary for operation on the air, because The output power of a modified receiver or transceiver attachment for it, as a rule, does not exceed 2-3 watts. In this case, it is most advisable to use a hybrid power amplifier (PA), which allows you to obtain a power gain of up to several hundred.

Some radio amateurs are distrustful of hybrid RAs, believing that such amplifiers do not provide high-quality signals. In fact, hybrid RAs provide high-quality signals that are in no way inferior to amplifiers made using classical circuitry. It should be noted that hybrid amplifiers require careful adjustment and understanding of the processes that occur.

There are publications of hybrid RAs using both bipolar and field-effect transistors; unfortunately, both of them have disadvantages, which I will briefly discuss.

The main disadvantage of bipolar transistors is the need to set a large initial current of 100 mA or more to bring the transistor to the beginning of the linear section of the characteristic. A large initial current of the transistor and, accordingly, the lamp, reduces the efficiency of the amplifier and leads to overheating of the lamp anode even in the absence of an excitation signal. A small initial current leads to signal limitation from below and noticeable nonlinear distortions.

The disadvantage of field-effect transistors is the high residual voltage at the drain (8...12 V) and, accordingly, high internal resistance. The current of a field-effect transistor, for example KP901, begins to be limited at about 300 mA. Since after reaching the specified current, an increase in the excitation amplitude does not lead to an increase in the drain current, the signal is limited from above.

The proposed hybrid RA uses a bipolar transistor. The inherent disadvantages of this option are eliminated using special circuitry, which allows you to separately set the initial current of the lamp and transistor, for example: the lamp current is 15 mA, and the transistor current is 120 mA.

The amplifier operates two 6P45S tubes with a KT922B transistor in the cathode. Unlike known circuits, voltage from a current stabilizer made on transistors VT5 and VT6 is supplied to the collector of transistor VT4 through decoupling choke L7 and protective diode VD11. Through the transistor VT4 in the cathode of the lamps, the total current of the lamps VL1 and VL2 and the stabilizer on VT5 and VT6 flows. Each of these currents is independently adjustable and can be set to a given value, thereby ensuring the required operating mode for both the lamps and the transistor. The current passing through the lamps and transistor VT4, in the absence of excitation voltage, is the initial current of the lamps. When the excitation voltage is applied, the current through the lamps and transistor changes and is proportional to the excitation level. That part of the current that flows to transistor VT4 from the stabilizer is always constant and does not depend on the excitation level. A chain of two diodes VD7, VD8 and a zener diode VD6 protects the transistor VT4 from overvoltage. The filament voltage for the lamps is supplied through inductor L6, which eliminates the harmful influence of the capacitance between the cathode and the filament. The excitation voltage is supplied to the base of transistor VT4 through a broadband step-down transformer T1, which matches the 50-ohm PA input with the low-impedance input of the transistor. The ALC voltage is removed from the emitter of transistor VT4 and adjusted using potentiometer R25.

The node on the DD1 chip allows you to switch the PA to transmission mode. The control procedure is as follows: after closing the pedal contact to the body, the key on VT1 locks RX; after a given time interval, antenna relay K1 connects the antenna to the PA; and finally, after a time delay, the transmission mode is established using relay K2. After releasing the pedal, the process is reversed: TX is turned off; The antenna switches and the receiver is allowed to operate.

Establishing RA begins with setting a current of 100-110 mA in the current stabilizer at VT5, VT6. To adjust the stabilizer, it is necessary to disconnect the collector of the VT5 transistor from the rest of the circuit and connect it through a milliammeter and a 300 Ohm resistor connected in series with it to the housing. The stabilizer current is set by resistor R27, the value of which is determined by the formula R = 0.625 / I, where the resistance is in Ohms, the current is in Amperes. In our case, a 6.25 Ohm resistor is required. There is no standard resistor of this value, so you should connect two resistors 6.8 Ohm and 68...82 Ohm in parallel. Next, after restoring the current stabilizer circuit, by adjusting potentiometer R14, the initial lamp current is set to 15...20 mA (PA - in transmission mode, excitation is not applied). If the initial current does not fall within the specified limits, it is necessary to change the value of resistor R11. The total current through transistor VT4 must be equal to the sum of the currents through the lamps and the current stabilizer. The base current of transistor VT4 is small and may not be taken into account. Current control VT4 is carried out by the voltage drop across resistor R20.

The last stage is setting up the PA contour system. The starting point for tuning is the anode current of the lamps with excitation applied and the anode circuit detuned.

When adjusting the excitation level, it is necessary to set the anode current of the lamps, with a detuned circuit - 620 mA. This operation must be performed very quickly, because... in this case, all the supplied power is dissipated on the anodes of the lamps, and they may fail. Now, by adjusting the antenna capacitor and adjusting the anode capacitor of the loop system, until the anode current declines, set the latter at 550...560 mA. The decay of the anode current at resonance, in relation to the “drive” current, should be 10%; it is this value of the decline of the anode current that ensures good linearity and high efficiency of the RA in SSB mode. In the CW mode, the decline in the anode current can be 20%, in this case the maximum PA power is achieved and the thermal conditions of the lamps are facilitated. It should be especially emphasized that when setting up the anode circuit, the excitation signal must be either single-tone or CW. The use of a two-tone signal or voice when setting up the PA, as well as the use of various field strength indicators, does not allow the amplifier to be properly configured and leads to the appearance of intermodulation distortion and, as a consequence, to an expansion of the emitted frequency band.

The proposed amplifier, with a high-quality circuit system, provides a peak power in SSB mode of 385 watts, with an efficiency of 68%, and the level of intermodulation distortion does not exceed -30 dB. The input voltage required to achieve maximum power does not exceed 10 V into a 50 Ohm load.

A few general notes. 6P45S lamps have anodes located not quite symmetrically relative to the grids, which leads to uneven heating of the anode and a decrease in the power dissipated by it. Therefore, maximum RA power can only be provided by specially selected lamps with uniform heating of the anode.

In the 6P45S lamp, the conductor connecting the anode to the anode cap inside the lamp is made of thin copper wire, which can melt when the RA operates with maximum power at the highest frequencies. Therefore, when operating on the 24 and 28 MHz bands, it is necessary to reduce the PA output power by 30%.

An amplifier using 6P45S lamps requires a fairly low load resistance and a correspondingly large variable anode capacitor. Since such capacitors are currently in very short supply, it makes sense to replace it with a set of constant capacitors, switched by a range switch. In this case, a ball variometer can be used as a loop inductance; it is also used to tune the anode circuit to resonance.

The proposed version of the loop system has a narrower range of matched resistances than in a conventional P. loop and requires the use of antennas with cable reduction.

And finally, about some design features of the RA.

Two holes with a diameter of 58 mm are cut into the chassis for installing lamps. Two lamp sockets are mounted on an aluminum plate located under the chassis so that the lamps are recessed by 18 mm after installation. The T5 transistor is installed on a 40x40 mm needle radiator.

It is recommended to lay a common body bus made of thin copper or foil PCB with a width of 15...20 mm between the body part of the antenna connector and the lamp sockets. All blocking capacitors connected to the lamps, as well as all parts of the circuit system that must be connected to the housing, must be connected to the housing bus. There is no need to isolate the chassis bus from the chassis.

Literature:

1. Zhalnerauskas V. Hybrid linear power amplifier. "Radio" No. 4 1968
2. Andryushchenko B. HF amplifier “Retro”. “Radiomir HF and VHF” No. 4 2002

The amplifier is designed for linear amplification of single-sideband, telegraph and AM signals in the ranges of 10...80 m. When amplifying telegraph and AM signals (in carrier mode), the input power is 200 W, when amplifying single-sideband signals, the average input power (when pronouncing a long “a” in front of the microphone) is also 200 W, while the peak input power can reach 400-500 W. The amplifier efficiency is 65-70% depending on the operating range. The amplifier uses four G811 lamps connected in parallel according to the OS circuit (Fig. 1).

A. Jankowski (SP3PJ)
Despite the general trend of using semiconductor devices in all technical devices, it is still necessary not to forget that tube HF power amplifiers (with an output power of more than 100 W) are much simpler to manufacture and more stable in operation. Experimenting with transistor devices is an expensive pleasure, because as someone aptly noted, no one dies so quietly, so quickly and as surely as a transistor.

Who needs power amplifiers? Few amateurs work QRP, but most sooner or later begin to dream of increasing the transmitter power. However, you must be aware that in order for the correspondent to notice a change in signal strength of one S scale point (6 dB), the output power of the transmitter must be increased fourfold, and it does not matter whether it is a local connection or a QSO with DX.

Vyacheslav Fedorchenko (RZ3TI), Dzerzhinsk, Nizhny Novgorod region. Many radio amateurs construct short-wave power amplifiers using direct filament lamps, such as GU-13, GK-71, GU-81. These lamps are inexpensive, easy to use, have a highly linear characteristic and do not require forced cooling. The main positive quality of these lamps is their readiness for work within one or two seconds after power is applied. According to the proposed description, more than a dozen structures were manufactured, which showed excellent technical characteristics, good repeatability, ease of setup and operation. The design is designed to be repeated by averagely qualified radio amateurs.

V. Gnidin UR8UM (ex,UR4UAS) I took the amplifier circuit from the article by V. Drogan (UY0UY) as a basis. “HF power amplifiers” I simplified the circuit a little, adapting it to fit the parts I have, so to speak, into a budget option. I present for review what happened.

Oleg Platonov (RA9FMN), Perm
This amplifier operates on the amateur bands 3.5-28 MHz. With an input signal power of 25...30 W, its output power in SSB mode on the 3.5-21 MHz bands will be at least 600 W and at least 500 W on the 24 and 28 MHz bands. The input impedance of the amplifier is 50 Ohms.

It is made on two GMI-11 pulse generator tetrodes, connected in parallel according to a circuit with a common cathode

Using a hybrid amplification circuit and impedance matching with an input P-circuit, we pump the signal up to a power of 150-160 W at an anode current of two GU-50s - about 300 mA in key-press mode. It is also advisable to control the current of the screen grids and not exceed its value more than 40 mA for two lamps. 250V x 0.02A = 5W - the maximum permissible level of power dissipation on the screen grid for one lamp. A protective diode will protect the stabilizer transistor in the event of a possible lamp shooting through the grid.

Typically, a power amplifier for a radio station or HF transceiver is built on "GU..." type lamps or on powerful high-frequency transistors. Both of these options may not always be acceptable. GU series lamps are relatively scarce, and powerful RF transistors, although they can be purchased, are prohibitively expensive. In addition, to build an output stage with a power of more than 100 W, several such transistors will be required, plus labor-intensive high-frequency transformers. The power amplifier described in this article is built according to a hybrid circuit using two relatively affordable transistors (KT610A and KT922V) of medium power, and one 6P45S lamp, which was widely used in the horizontal scanning output stages of tube TVs and, in this regard, is also relatively accessible and cheap.

I. AUGUSTOVSKY (RV3LE), Smolensk region, Gagarin The idea of building a push-pull amplifier using electronic tubes is not new, and the circuitry of this amplifier, in principle, is no different from the circuitry of building push-pull amplifiers using transistors. It should be noted that current lamps work best in this circuit, i.e. lamps with low internal resistance, which are capable of providing a significant pulse of anode current at a low supply voltage. These are lamps of the 6P42S, 6P44S and 6P45S types. However, I was able to build an amplifier with good characteristics using a GU-29 type lamp.

Hybrid power amplifier circuits. Hybrid linear HF power amplifier

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